Patch Loaded Slot Antenna for Super Wideband Applications with Dual-Band Notch Characteristic

This paper presents a patch loaded slot antenna for super wideband (SWB) application. To obtain SWB characteristic, the proposed antenna geometry combined a rectangular slot and an overlying patch excited by a trident shaped microstrip feed. It is observed that the hybrid nature of the proposed antenna effectively enhances the impedance bandwidth up to 120%, by combining the resonance of both patch and slot. Besides, it is investigated that after converting the conventional tapered feed into the trident shape feed, the impedance bandwidth is increased further from 120 to 167% ranging between 1.25 and 15 GHz. Moreover, one U-shaped slot and two L-shaped stubs are inserted into the antenna design to introduce the dual-band rejection property from 1.8 to 2.4 GHz (GSM 1800, Wi-Fi 2.1 and 2.4) and 3.1–4.2 GHz (WiMAX and C-band). Further, to validate the simulation results a prototype is practically constructed and tested. The measured result demonstrates that the designed antenna offered an impedance bandwidth of 170.3% from 1.2 to 15 GHz. Both frequency and time domain analysis ensure the suitability of the designed antenna for SWB applications.


Introduction
Revolutionary progression in wireless technology builds a huge demand for the super wideband (SWB) antenna to support a very high data transmission rate. The bandwidth ratio of f h /f l (where f h is highest and f l is lowest frequency edge of the frequency response) is generally used to differentiate the SWB antenna from the other readily available wideband antenna or ultra-wideband (UWB) antenna. Practically, antenna offering f h /f l ratio more than 10:1 is considered as SWB antenna [1].
Unlike UWB, no specific frequency band is allotted for SWB technology. It has an extremely wide bandwidth, which can completely cover many frequency bands like GSM 900, 1800, Wi-Fi, and entire UWB simultaneously.
However, some narrow frequency bands co-located within UWB and SWB range like WiMAX (3.3-3.6 GHz), downlink of C-band (3.7-4.2), Wi-Fi 2.1and 2.4 GHz may degrade the performance of SWB antenna by causing undesired electromagnetic interference from these coexisting bands. One way to reduce their interference is to connect several band stop filters with SWB antennas; however, it will increase the system complexity and size tremendously. Another method is to design the SWB antenna with band rejection features without affecting the size and complexity. In literature, different types of SWB antennas, including fractal, printed monopole has been proposed [2][3][4][5][6][7]]. However, none of them neither covering important frequency bands like GSM-900, 1800, Wi-Fi 2.1 and 2.4 GHz nor offering band rejection property. In order to cover lower frequency applications with broadband behavior, several SWB antenna has been proposed in the recent past [8][9][10][11][12][13][14][15][16]. Although, the reported works covered most of the lower frequency band, but not ensured band notch characteristics.
Very few of SWB antennas offered single [17][18][19][20] dual [21,22], and triple [23] band notch characteristics. However, none of the design focused on band rejection outside the UWB range. For instant, a monopole SWB antenna with a single-band notch is reported in [18]. Although the proposed antenna is compact and operated under low frequency region 0.9 to 100 GHz, nonetheless the suppressed band is only one which falls outside the UWB band. Similarly, a SWB monopole antenna with tri-band notch characteristic is reported in [23], but all the rejection bands belong to UWB range. Moreover, the antenna do not support the lower frequency region. In this study, a hybrid SWB antenna operating between 1.2 and 15 GHz with dual-band notch property is proposed. In order to achieve broad impedance bandwidth, a trident shape feed line is used to excite the antenna. It shifts the upper edge of frequency response from 5.1 to 15 GHz and enhance the bandwidth from 120 to 167%. Further, to obtain the dual-band rejection inside (entire WiMAX frequency band and downlink frequency of C-band) and (GSM-1800, Wi-Fi 2.1, 2.4 GHz) outside of the UWB range, one U-shape slot and two L-shape stubs are introduced in patch and slot, respectively. A comprehensive investigation is done in the subsequent section to achieve the broad impedance bandwidth with strong band rejection characteristics. Figure 1 illustrates the geometry of the designed SWB antenna. It is designed on an inexpensive substrate FR-4 having dielectric constant of 4.3 and height 1.5 mm. As seen, the bottom layer of the substrate consists of a rectangular slot, whereas; the top layer has a rectangular patch. The feeding mechanism of the antenna adopts a trident shaped microstrip line comprising three arms of different sizes. The central arm is the linearly tapered 50Ω transmission line connected to two inverted L-shaped stubs at either side, which comprises the second and third arm of the feed. The proposed antenna is primarily designed for broad impedance bandwidth afterward, a rectangular U-shaped slot and two L-shaped stubs are introduced in patch and star shaped slot to achieve dual-band rejection property.

Antenna Design and Parametric Study
In the proposed antenna, the broadband characteristic is achieved in two simple steps. First, conversion of slot antenna into hybrid antenna by embedding a rectangular patch of size P l ×P w at the upper portion of the substrate, as shown in step 2 of Fig. 2. The added rectangular patch introduced a new resonance that gets merged with the slot resonance after optimizing the dimension and position of both slot and patch. Subsequently, two inverted L shaped stubs defined by f h ×f l are attached with feed line near to the bottom edge of the patch. This yields a strategic perturbation of magnetic current distribution in both slot and patch, which enormously enhances the impedance bandwidth. The optimized values of different design parameters of the antenna are listed in Table 1.  Fig. 2. It can be noticed that the antenna radiates in high frequency region and offers very low impedance bandwidth when it consists of a rectangular slot excited by the microstrip feed. However, after introducing a rectangular patch at the other side of the substrate, it achieved an impedance bandwidth of 120%, which is approximately 12 times larger than the slot antenna but radiates mainly in the low frequency region. Afterward, the conventional feed is converted into a trident shaped feed, which results in further improvement in impedance bandwidth from 120 to 167% by shifting the upper edge of resonance from 5.1 to 15 GHz. The gain comparison corresponds to different design steps is illustrated in Fig. 3b. It can be seen after embedding the patch to the slot and converting the conventional feed to trident shaped feed, the antenna gain is drastically improved in lower and upper frequency regions, which was contradictory otherwise in steps 1 & 2. The maximum gain variation is observed at 2 GHz, where the antenna gain is increased from -10 dBi to 4.16 dBi i.e., total 141% improvement.
Though the additional stubs at feed line end contribute to the bandwidth enhancement from 5.1 to 15 GHz, therefore variation of their dimensions strongly affects the impedance matching in this frequency range. Fig. 4a and b demonstrate the simulated VSWR for different values of stubs length (f l ) and width (f w ), respectively. As seen in Fig. 4a, varying the f l from 2.5 mm to either 1.5 mm or 3.5 mm, the impedance matching of the antenna is affected. A similar effect can be observed when the stub width (f w ) is reduced from 5.5 mm to 3.5 mm.
By lowering the f w , the impedance matching gets worse, especially between 8 GHz to 11 GHz. As seen in Fig. 4b, adjusting the stub width, the band notch characteristic of the antenna can be controlled. Fig. 5 illustrates the simulated VSWR against the different offset value between the bottom edge of ground slot and patch. It can be seen that a small shift in offset value made a drastic change in antenna performance. The optimum impedance bandwidth is achieved for an offset value of h=4 mm. It could also be observed that by connecting or disconnecting the  . 3 a Simulated S11 against different design steps and b simulated gain against different design steps patch (or using switch between feed and patch), the matching of either lower region or upper region could be controlled. This way a bandwidth re-configurability can be achieved in the proposed circuit. Besides, broad bandwidth, our aim is to generate dual-band notch characteristics in the proposed SWB antenna. For this one U-shaped slot and two L-shaped stubs are introduced in the patch and slot, respectively. The addition of slot and stub to the proposed antenna changes the impedance bandwidth slightly by downshifts the lower edge frequency from 1.3 to 1.25 GHz. The optimized dimension and position of U-shaped slot and L-shaped stubs produce the strong current distribution in the opposite direction of the patch and slot, respectively, which results in sharp band rejection from 1.8 to 2.4 GHz and 3.1 to 4.2 GHz. To obtain the band rejection at WiMAX and lower frequency region of C-band, a U-shape slot is etched at the lower portion of the patch, as shown in Fig. 1. As seen that there are two major parameters slot length u l and slot width u w , which can directly control the attenuation level strength and position of the rejection band. The rejection band frequency can be calculated by using the equation1.
(1)  where c is speed of light and L total = u w +2u l and ϵ eff = (ϵ r +1)/2. The simulated VSWR corresponds to different values of u l and u w are depicted in Fig. 6a and b, respectively. As seen, increasing either slot length or width, the center frequency of the rejection band is shifted downwards which validates the equation 1. Moreover, increasing the slot width, the rejection band gets wider and stronger. It can be noticed that the lower edge of the rejection band is greatly influenced by slot width u w , whereas u l curbs the movement of the entire rejection band. Apart from slot width and length, the offset position of slot may also contribute to the rejection band moderation. Fig. 7 illustrates the simulated VSWR against the different slot offset positions. As seen, the upper edge of the notch band is shifted towards the higher frequency by varying the offset value (u h ) from 0.5 mm to 2 mm. The optimum value of U-shaped slot length, width and position, where the proposed antenna obtains the appropriate band rejection are u l =9, u w = 9 mm and u h =0.5 mm.
In order to suppress GSM-1800, Wi-Fi 2.1 and 2.4 GHz, two L-shaped stubs are attached at either side of the rectangular slot, as shown in Fig. 1. Equation 2 shows the relationship between the notch frequency and the stub length. Here, it can be noticed that increasing either S w1 or S w2 , the notch frequency will reduce which can be confirm from Fig. 8a and b.  As seen, increasing the stub length in both X and Y direction, the center frequency of the rejection band is shifted downwards. Moreover, increasing the stub length the rejection band becomes broader and stronger. Here, the upper edge of the rejection band is almost unaffected, whereas lower shifted downwards. Fig. 9, illustrates the effect of offset (S h ) value on antenna performance. It can be seen that the small movement of stub in upward or downward direction does not much affect the antenna performance.
To support the above results, electric current distribution at the center frequency of both rejection bands is shown in Fig. 10. As seen, the maximum current is distributed along the inverted L-shaped stubs at 2.1 GHz (center frequency of the first rejection band). As seen in supporting figure, the electric current changes its direction at the connecting point of the stub and becomes completely opposite to the direction of ground plane current. As a consequence, this stub cancels out the antenna radiation in rejection band. Here, stubs act as a resonator and their effective length is nearly equals to quarter wavelength calculated at 1.8 GHz. At 3.6 GHz and 4.1 GHz, the current is mainly concentrated at the periphery of the U-shaped slot. As seen, the current is in upward direction at the interior edge of the Fig. 8 Simulated VSWR for different value of stub a horizontal length S w1 and b vertical length S w2 Fig. 9 Simulated VSWR for different value of stub offset position S h U-slot whereas, at the exterior edge, it is in the downward direction. Therefore, destructive interferences occur in the above frequency range and result in no radiation. The simulated and measured VSWR of the designed antenna is illustrated in Fig. 11. As seen, both the results are in well agreement and notches are perfectly coming at desired frequency bands. Further, at higher frequencies small ripples can be observed that may be due to cable and connector losses.
The measured and simulated radiation pattern of the designed antenna at five different The measured and simulated radiation pattern of the designed antenna at five different frequencies is illustrated in Fig. 12. As seen, in E-plane antenna ensures almost monopole like pattern, whereas in H-plane it exhibits Omni-directional pattern throughout the frequency range. Above 10 GHz, the antenna shows Omni-directional pattern in both planes. From the above results, we can conclude that the proposed antenna offers approximately stable radiation pattern throughout the operating band. Figure 13a and b, shows the simulated gain and radiation efficiency with and without the band rejection. From Fig. 13a, it can be seen that the antenna attains almost stable gain throughout the frequency range. It is varying between 3.3 and 5.5 dBi (except at the rejection band) and exhibits a maximum gain of 7.15 dBi at 6.4 GHz. Though, the antenna operating over a large frequency range, therefore, this gain variation could be acceptable. Further, at both notch frequencies, gain and efficiency are reduced considerably. At first notch frequency, the gain is reduced from 4.1 to −2.3 dBi, whereas at the second notch, it drops from 5.4 to −6.4 dBi. In addition, by activating/deactivating (through switch) two stubs in the slot and U-slot in the patch reconfigurable band rejection capability could also be realized in the same circuit. The performance comparison of the proposed antenna with other similar reported works is shown in Table 2. It can be observed that compared to earlier reported SWB antennas, the proposed antenna offers the wide impedance bandwidth with lower frequency convergence and dual band notch property.

Time Domain Analysis
In order to check the reliability aspect of the proposed SWB antenna the time domain analysis is as important as frequency domain analysis. To execute it, two same antennas have been realized and placed beyond far-field region at a distance of 30cm in two different configurations: (1) face-to-face and (2) side-by-side as shown in Fig. 14. The CST Microwave Studio is utilized to perform the time domain analysis. A short duration Gaussian pulse is utilized to transmit and receive the signal in both the cases. Fig. 15a illustrates the normalized amplitude of the input and received pulse for both side-by-side and face-toface configurations. The correlation between the input and output pulses provides the level of distortion in the shape of transmitted signal. Ideally in case of ultra or super wideband antennas the shape of received pulse should be an exact replica of transmitted pulse.
To maintain the pulse shape the group delay should be constant or phase should be linear over the entire pass band region of SWB antenna. Fig. 15b shows the group delay variation over frequency in both cases. It can be seen that in both arrangements, the proposed antenna reveals a reasonably constant group delay in the entire frequency range except in band notch region. This indicates the phase linear characteristic of the proposed antenna in the far field region. Fig. 16 demonstrates the |S21| magnitude variation for both arrangements. In the entire frequency range isolation is more than −30 dB in side-by-side case, whereas in face-to-face configuration it is more than −20 dB. The higher value of isolation |S21| in side-to-side arrangement signifies that the correlation between transmitted and received pulse is less compared to face-to-face arrangement.